Zero Voltage Soft Switching Scheme for Power Converters

ABSTRACT

A control scheme and architecture for a power conversion circuit employs two bidirectional switches and a zero voltage switching (ZVS) scheme for the high-side switch. Methods of incorporating the control scheme into multiple power conversion circuit topologies are disclosed. Methods of device integration including co-packaging and monolithic fabrication are also disclosed.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims priority to Provisional Application No.62/059,008, filed Oct. 2, 2014, titled “ZERO VOLTAGE SOFT SWITCHINGSCHEME FOR POWER CONVERTERS”, which is hereby incorporated by referencein its entirety.

FIELD

The present invention relates generally to soft switched powerconverters, and in particular to zero voltage soft switching (ZVS) inFlyback, Forward, Buck, Buck/Boost, Boost and other topologies.

BACKGROUND

Switched mode power converters are ubiquitous and are often used toconvert one form of power to another. For example a Flyback convertermay be employed in an electronic system to convert a high voltagealternating current (AC) bus (e.g., 220 volts AC) to a low voltagedirect current (DC) bus (e.g., 5 volts DC) that may power a componentsuch as a cellular phone. Switched mode power converters have threebasic figures of merit: cost, size and efficiency. To be accepted inlarge volume applications, power converters must meet minimumrequirements for all three specifications.

Power loss in a switch, or field-effect transistor (FET) employed inswitched mode power converters comes from two sources. FETs have aresistive element, that dissipates power as current is conducted throughthe device. The resistive parameter is typically called “on-resistance”,or RDS(ON) (i.e., resistance from drain to source when the FET is biasedon). These conduction losses are inversely proportional to the size ofthe FET (i.e., the larger the FET, the lower its RDS(ON) and, therefore,the lower its conduction loss). The other source of power loss isthrough switching losses. Every time a solid-state switch is turned onor off there is energy loss, as described in more detail below.

Increased switching frequency has been a significant factor in theimprovement in the cost and size of switched mode power converters.Increased switching frequency typically reduces the size of peripheralcomponents and provides improved transient response for demandingapplications. However, as discussed above, increased switching frequencyresults in increased power loss and decreased efficiency for the powerconverter.

Two major factors contribute to power loss from switching thetransistors: turn-on loss, or the energy used to charge drain-sourcecapacitance (also commonly referred to as output capacitance or Coss);and crossover loss, or the energy lost during turn-on and turn-offtransitions (i.e., the current and voltage overlap area as the switchtransitions between states).

With regard to output capacitance, or Coss, as the FET switches on andoff with a voltage potential across it, its intrinsic parasiticcapacitance stores and then dissipates energy during each switchingtransition. Essentially there is an embedded capacitor within the switchthat must be charged and discharged with each switching cycle. Theoutput capacitance losses are proportional to the voltage across theswitch, the switching frequency and the value of the parasiticcapacitance. As the physical size of the FET increases, its outputcapacitance also increases. Therefore, as discussed above, increasingFET size may reduce RDS(ON), however it also increases outputcapacitance and thus increases switching loss.

To reduce high frequency switching losses, ZVS schemes have beenproposed using silicon power devices, where the voltage potential acrossthe switch is reduced to near 0 volts prior to operating the switch.This can significantly reduce crossover loss as there is almost novoltage potential across the switch when the switch is operated. Schemeshave also been proposed that discharge the output capacitance (Coss) ofthe switch and recycle the energy back into the system, significantlyreducing losses due to output capacitance, or Coss. However, ZVS schemeshave not been widely adopted, especially in Flyback applications, asthey are too costly and intractable.

For example, silicon devices have a relatively large output capacitance(Coss) (e.g., 200 picofarads) which takes a relatively long amount oftime to charge. In high frequency applications, the time required tocharge the output capacitance may limit the switching frequency of theconverter. Further, silicon devices switch relatively slowly, (e.g., onthe order of 20 nanoseconds) which also limits the switching frequency.Yet further, silicon devices are vertical structures typicallyfabricated such that the substrate is a drain terminal. Thus they do notlend themselves easily to monolithic integration with other devices asthe other devices would be fabricated on the drain connection. Thissignificantly restricts packaging and integration options to savepackaging cost and size. Thus, in a two switch silicon-based powerconverter each switch is typically a separate device. The switch driverand controller circuits are also typically separate devices furtherincreasing costs and increasing the driver delay due to packagingparasitics. Moreover, especially for high voltage applications (i.e.,greater than 100 volts), silicon devices have poor performancecharacteristics and require large, slow, expensive driver circuits tooperate. These and other factors have limited the adoption of ZVSarchitectures for silicon-based high frequency, high voltageapplications.

SUMMARY

In one embodiment a new control scheme that drives bidirectional switchconverters, such as a two-switch Flyback, using ZVS is disclosed. In oneembodiment the new control scheme achieves ZVS, while minimizingexcessive ripple current loss and maintains compatibility with pulseskipping or pulse frequency modulation (PFM) controller modes.

In one embodiment a power conversion circuit comprises a firstsolid-state bidirectional switch connected between a first terminal of avoltage source and a switch node. A second solid-state bidirectionalswitch is connected between the switch node and a second terminal of thevoltage source. The second switch is configured to turn on before thefirst switch with a duration that is less than a time the first switchis off.

In one embodiment the power conversion circuit is configured to operatein a discontinuous mode while in another embodiment the circuit isconfigured to operate in a continuous current mode. In furtherembodiments the first and second solid-state bidirectional switches areGaN-based devices that operate between 50 kHz and 100 MHz.

In further embodiments the power conversion circuit is configured tooperate at voltages between 100 V and 600 V. In one embodiment, thefirst and second solid-state bidirectional switches may be co-packagedwhile in other embodiments a switch driver may be co-packaged with thefirst and second solid-state bidirectional switches. In yet furtherembodiments the first and second solid-state bidirectional switches maybe monolithically integrated on a first die comprising GaN. In otherembodiments the first switch driver circuit and the first solid-statebidirectional switch are monolithically integrated on a first die and asecond switch driver circuit and the second solid-state bidirectionalswitch are monolithically integrated on a second die.

In one embodiment the power conversion circuit may be disposed within aunitary electronic component. In some embodiments the component may bemanufactured from an overmolded printed circuit board, while in otherembodiments the component may comprise an overmolded lead frame. Infurther embodiments the component may comprise a driver circuitconfigured to turn on and off the first and second solid-statebidirectional switches. The switches may be disposed on a monolithicsemiconductor substrate comprising GaN.

In one embodiment, a power conversion circuit may include a firstsolid-state bidirectional switch having a first switch outputcapacitance and connected between a first terminal of a voltage sourceand a switch node. A second solid-state bidirectional switch isconnected between the switch node and a second terminal of the voltagesource. The second switch is configured to turn on before the firstswitch with a duration that is less than the time that the first switchis off and to remain off for duration that is adequate to allow thefirst switch's output capacitance to discharge to approximately 0 voltsbefore the first switch is turned on. In further embodiments a capacitoris connected in series between the second switch and the second terminalof the voltage source and is configured to reverse a direction ofcurrent. In one embodiment a transformer is connected in series betweenthe switch node and the second terminal of the voltage source. Thecircuit may further be configured to turn on the first switch such thatcurrent flows through the transformer, and subsequently turn off thefirst switch such that the currently decreases to 0 amps and charges thecapacitor. Next, the second switch is turned on allowing the capacitorto reverse the flow of current through the transformer. The secondswitch may then be turned off such that the reverse current dischargesthe output capacitance of the first switch to approximately 0 volts. Thefirst switch is then turned on and the process is repeated.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit schematic of a Flyback converter using twobidirectional switches according to an embodiment of the invention;

FIG. 2 is a plot of the switch node voltage for the circuit illustratedin FIG. 1;

FIG. 3 is a plot of the gate voltages for the MFET and the SFET in thecircuit illustrated in FIG. 1;

FIG. 4 is a plot of the current in the MFET and the SFET in the circuitillustrated in FIG. 1;

FIG. 5 is a blow up of a portion of the switch node voltage plotillustrated in FIG. 2;

FIG. 6 is a blow up of a portion of the gate voltage plot illustrated inFIG. 3;

FIG. 7 is a blow up of a portion of the current in the MFET and the SFETplot illustrated in FIG. 4;

FIG. 8 is waveform of a converter operating in DCM according to anembodiment of the invention;

FIG. 9 is waveform of a converter operating in CCM according to anembodiment of the invention;

FIG. 10 is an example of pulse frequency modulation (PFM) that may beused according to an embodiment of the invention;

FIG. 11 is an example of different power conversion topologies accordingto embodiments of the invention;

FIGS. 12A-12C are different DC to DC power converter topologiesaccording to embodiments of the invention;

FIG. 13 is an example of waveforms for a DC to DC Buck converteraccording to an embodiment of the invention;

FIG. 14 is an example of co-packaged devices according to an embodimentof the invention;

FIG. 15 is an example of co-packaged devices according to an embodimentof the invention;

FIG. 16 is an example of co-packaged devices according to an embodimentof the invention;

FIG. 17 is an example of co-packaged devices according to an embodimentof the invention;

FIG. 18 is an example of monolithically integrated devices according toan embodiment of the invention;

FIG. 19 is an example of a two switch half-bridge Flyback circuitaccording to an embodiment of the invention;

FIG. 20 is an example of a monolithically integrated GaN transistorincluding its own drive circuit according to an embodiment of theinvention; and

FIG. 21 is an example of a monolithically integrated GaN transistorincluding its own drive circuit according to an embodiment of theinvention.

DETAILED DESCRIPTION

Certain embodiments of the present invention relate to power conversioncircuits. While the present invention can be useful for a wide varietyof power conversion circuits, some embodiments of the invention areparticularly useful for AC to DC and DC to DC converters that use ZVSbidirectional switches, as described in more detail below.

Many electronic devices such as smart-phones, media players, and tabletcomputers require low-voltage DC power to operate. Some electronicdevices may be configured to connect to an AC mains to receivehigh-voltage AC power. To make the AC power useful to the electronicdevice, it is typically rectified to a high DC voltage which maysubsequently be converted to a lower DC voltage by a DC to DC powerconversion circuit. In some embodiments a ZVS high-efficiency DC to DCpower conversion circuit that employs bidirectional switches and a novelcontrol scheme may be used.

As defined herein, a bidirectional switch enables the conduction ofcurrent in two directions when in an on state and prevents theconduction of current in any direction when in an off state. It canfurther be defined as a switch that doesn't have an antiparallel bodydiode and it may block an unequal amount of voltage in either direction(i.e., it can block more voltage in one direction than the other). Insome embodiments, a bidirectional switch may be fabricated on a GaNsubstrate. The lack of an antiparallel body diode in a bidirectionalswitch avoids reverse recovery problems experienced in silicon-basedMOSFETs. This enables the bidirectional switch to change statesrelatively fast and to charge a Flyback clamp capacitor, which may beused to energize a transformer to achieve ZVS, as discussed in moredetail below.

Comparatively, a silicon-based MOSFET has an antiparallel body diode,which is a PN junction diode that lies inside of the MOSFET between then-region in the drain and the P-well in the source. The PN junctionantiparallel body diode has reverse recovery charge, which causes theMOSFET to be relatively slow turning off such that it can block voltage.

Now referring to FIG. 1, a non-limiting example of a two-switch Flybackcircuit 100 of one embodiment is illustrated. In some embodiments a mainfield-effect transistor (MFET) 105 and a second field-effect transistor(SFET) 106 may be used as bidirectional switches in the circuit. Infurther embodiments a novel control scheme may be used by circuitcontroller 107 to command SFET driver 108 and MFET driver 109 such thatSFET 106 may be turned on slightly before the MFET, creating enoughcurrent to enable ZVS of MFET 105 (i.e., zero voltage soft-switching) asdiscussed in more detail below.

As used herein, ZVS means that the bidirectional semiconductor switchmay be turned on or off only when the voltage applied across the switchis at or near zero (i.e., zero voltage switching or ZVS) and when theoutput capacitance, or Coss, is at or near zero charge. Switching losses(i.e., turning a switch off while it is conducting current or turning aswitch on when it has a voltage potential across it) may be asignificant contributor to power loss in the system. The use of ZVS mayresult in reduced switching losses, increased frequency of operation andin some embodiments, reduced electromagnetic interference (EMI)generation.

Continuing to refer to circuit 100 in FIG. 1, controller 107 may havelogic and control functionality such that it can control the operationof MFET 105 and SFET 106 by sending one or more control signals to SFETdriver 108 and MFET driver 109. In response to the one or more controlsignals, SFET driver 108 and MFET driver 109 may then send one or moregate drive signals to MFET 105 and SFET 106. In response to the one ormore gate drive signals, MFET 105 and SFET 106 may then transitionbetween an on state and off state. In some embodiments, circuit 100 mayreceive power from an AC mains 115. A full-wave rectifier 110 mayconvert AC mains 115 to a DC source having cyclical voltage variationsat approximately twice the frequency of the AC mains. A smoothingcapacitor (Cs) 117 may be used to smooth the cyclical voltagevariations, creating a relatively steady high voltage DC source. MFET105 may be used to switch on and off the high voltage DC source. In someembodiments MFET 105 may be a bidirectional N-channel gallium-nitride(GaN) high-electron mobility transistor (HEMT), however in otherembodiments other switches may be used, as discussed in more detailbelow.

Now referring to FIGS. 2-7, plots of voltages and current within circuit100 are shown. FIG. 2 illustrates the switch node (SW) 120 (see FIG. 1)voltage (Vsw). FIG. 3 illustrates gate voltages of MFET 105 and SFET106. FIG. 4 illustrates the current in MFET 105 and SFET 106. FIGS. 5-7are expansions of FIGS. 2-4 in the time range starting at 9.7microseconds and ending at 10.7 microseconds.

Now referring to FIGS. 2-4, at time t1 controller 107 applies voltage tothe gate of MFET 105, opening its channel and allowing current to flowfrom its drain (Dm) to its source (Sm). As shown in FIG. 2, the voltageat switch node V(sw) 120 is clamped to 0 volts and current flows indirection I1 illustrated in FIG. 1. In FIG. 4 the current within MFET105 is shown which linearly increases with respect to time up toapproximately 2.5 Amps. Transformer 125 will produce an opposing voltageacross its terminals in response to the changing current. The rate ofincrease of current through MFET 105 may be dictated by the parametersof transformer 125 as current builds within its windings. During thistime, transformer 125 is storing energy in the form of a magnetic field.Current in transformer 125 builds until MFET 105 is turned off at t2(see FIGS. 2-4).

When MFET 105 is turned off at t2 by controller 107, the voltagepotential from the full-wave rectifier will be removed from transformer125. As illustrated in FIG. 4, since MFET 105 is turned off the MFETcurrent rapidly transitions to 0 Amps and the voltage at the switch nodegoes back to the line voltage plus reflected output voltage,approximately 400 volts.

Transformers 125 resist changes in current so the stored energy in themagnetic field of the leakage inductance of the transformer willdischarge, maintaining the flow of current in direction I1 which nowflows through SFET 106. This current flows into and charges capacitor Cx130. Because the energy flows into capacitor Cx 130, it is conserved,improving the efficiency of circuit 100.

At t3, the current stops flowing when transformer 125 has exhausted theenergy stored within its leakage inductance. After t3, switch nodevoltage Vsw 120 (see graph shown in FIG. 2) may oscillate due toresonances within circuit 100. The oscillation between t3 and t4 mayhave a relatively high frequency and small amplitude, as secondary sidetransformer current still flows through the rectifier diode. In someembodiments this portion of the oscillation may be due to the leakageinductance of transformer 125 reacting with the output capacitance ofMFET 105.

After t4 the current in the secondary side of transformer 125 reduces tozero and no more current flows through the rectifier diode. The largeramplitude and lower frequency may be due to the magnetizing inductanceof transformer 125 interacting with the output capacitance of MFET 105.

Now referring to FIG. 2, controller 107 may be configured to monitor thevoltage oscillations of switch node (SW) 120 and at a peak voltage(i.e., when the switch node voltage is near 400 volts) the controllerturns SFET 106 on at t5. Stored charge in capacitor Cx 130 is thendischarged into transformer 125, creating current in direction 12 whichis in the opposite direction of the prior current, I1. Because SFET 106turns on at the oscillation peak, the step up voltage to get switch node120 voltage up to the line voltage is relatively small, resulting in afaster and more efficient transition. The step up and subsequentoperation of SFET 106 illustrated in FIGS. 2-4 is shown in greaterdetail in FIGS. 5-7. FIG. 7 shows a current spike at t5 when SFET 106 isturned on due to the voltage step up. After the step up, current beginsto build through SFET 106 and transformer 125 as illustrated in FIG. 7.

Capacitor Cx 130 continues to discharge and builds current (see FIG. 7)in transformer 125 and SFET 106 according to the characteristics of thetransformer. Current in transformer 125 builds until t6 when SFET 106 isturned off by controller 107. Current can no longer flow through SFET106 so it places a voltage potential on the output capacitance Coss ofMFET 105. The voltage potential draws current from the outputcapacitance Coss of MFET 105 causing it to discharge. In someembodiments only MFET 105 output capacitance may be discharged, howeverin other embodiments MFET 105 may conduct in a reverse direction afterCoss is fully discharged. The current flow discharges MFET 105 drain(Dm) voltage to approximately 0 volts at t7, as illustrated in FIG. 5.When switch node 120 voltage is at 0 volts the voltage potential acrossMFET 105 is approximately 0 volts and the output capacitance Coss of theMFET is discharged, enabling ZVS. The cycle starts over again at t1where MFET 105 is turned on by controller 107 and current builds intransformer 125.

In some embodiments it may be beneficial for the switches to be able towithstand high voltage potentials and/or to switch at high frequencies,particularly when the transmitter runs off AC mains. In one embodimentthe voltage potential across the switches may be in the range of 50-1000volts DC and in another embodiment in the range of 100-600 volts DC. Inone embodiment the voltage potential may be in the range of 100-250volts DC and in another embodiment it may be in the range of 250-600volts DC. In other embodiments the switching frequency may be in therange of 30 kHz-30 MHz while further embodiments it may operate in arange between 50 kHz-1 MHz. In another embodiment the switchingfrequency may be in the range of 100 kHz-500 kHz. In one embodiment theswitching frequency may be 100 kHz.

In some embodiments, one or more of switches 105, 106 may be a FET. Inone embodiment one or more of switches 105, 106 may be a GaNbidirectional FET. In another embodiment one or more of switches 105,106 may be a JFET, while in other embodiments it may be a different typeof FET or any other type of solid-state switch. GaN-based bidirectionalswitches may be particularly useful in embodiments that may be used toefficiently switch high voltage buses (e.g., 400 volts) at highfrequencies (e.g., 0.1-30 MHz) as described in more detail below. Insome embodiments the efficiency of the power conversion circuit may bein the range of 60% to 95%. In one embodiment the efficiency of thepower conversion circuit may be approximately 85%.

In some embodiments the power converter may be a single switch Flybackconverter and may operate in continuous conduction mode (CCM) ordiscontinuous conduction mode (DCM). In further embodiments CCM may beused at low input line and full load, and DCM may be used at high inputline and/or light load.

Now referring to FIG. 8, a waveform of an embodiment with a powerconverter operating in DCM is illustrated. In this embodiment theinductor current reduces to zero during each switch cycle. In furtherembodiments, this may occur at light loads and/or when the inductorripple is large.

Now referring to FIG. 9 a waveform of an embodiment with a powerconverter operating in CCM is illustrated. In one embodiment theinductor current never touches zero during the switching cycle. This mayoccur at high load and/or when the ripple current is small.

Now referring to FIG. 10, another embodiment is illustrated where pulsefrequency modulation (PFM) may be used to vary the switching frequencyto meet load demand and improve efficiency. In some embodiments, atlight load, the frequency may be reduced, and switching pulses may beskipped to reduce switching loss.

Now referring to FIG. 11, other embodiments may use different powerconversion topologies than a Flyback circuit. In this embodiment powerconversion circuit 1100 is a Forward converter. In one embodiment, therelationships of the SFET and the MFET may be similar to the Flybackembodiment discussed above. In this embodiment, the SFET turns onbriefly right before the MFET turns on to generate current in thetransformer. When the SFET turns off, the generated negative currentcreates ZVS for the MFET.

Now referring to FIGS. 12A-12C, embodiments may employ the novel ZVScontrol scheme discussed above in other DC to DC converter topologies.One of skill in the art will understand that such embodiments alsorequire control and FET driver circuits as discussed above. In oneembodiment the DC to DC converter may use inductors insteadtransformers. Further embodiments may have different circuitarchitectures and employ the novel ZVS control scheme discussed above.In such embodiments the relationships of the SFET and the MFET may besimilar to the Flyback embodiment discussed above. Further, the SFET maybe configured to turn on briefly right before the MEFT turns on togenerate current in the transformer. When the SFET turns off, thegenerated negative current creates ZVS for the MFET. For example, FIG.12A illustrates circuit 1200 which is an embodiment of a Buck converterthat may employ the novel ZVS control scheme. As another example, FIG.12B illustrates circuit 1210 which is an embodiment of a Boost converterthat may also employ the novel ZVS control scheme. As a further example,FIG. 12C illustrates circuit 1220 which is an embodiment of a Buck/Boostconverter that may further employ the novel ZVS control scheme. Othercircuit architectures may also employ the novel ZVS control scheme andare within the scope of this disclosure.

Now referring to FIG. 13 waveforms of an embodiment employing a DC to DCBuck converter architecture are illustrated. In one embodiment, the Buckconverter may operate in a discontinuous mode, where the inductorcurrent reduces to zero after the MFET is off (i.e., when the MFET'sgate is low). In further embodiments, the SFET can remain off when theMFET is off. In yet further embodiments, the SFET may turn on when theMFET is off and the inductor current is still positive. In otherembodiments it may turn off when the inductor current reduces to zero.In some embodiments this may be called synchronous rectification whichmay reduce conduction loss. In one embodiment, the SFET may be turned onright before the MFET turn-on event, generating a negative inductorcurrent. In yet further embodiments, after the SFET turns off, thenegative inductor may pull the Vds of the MFET to zero. When the MFETturns on at this moment, ZVS may be achieved for the MFET. In otherembodiments this concept may be applied to yet other topologies andcircuit configurations.

In further embodiments one or more of the features of the powerconversion circuit are:

-   -   1. There is main switch, and secondary switch. The second switch        turns on slightly ahead of the main switch to enable ZVS for the        main switch.    -   2. When the main switch has a voltage oscillation, a secondary        switch can turn on at the peak of the main switch voltage to        reduce any switching losses related to the second switch.    -   3. The second switch can turn on any time before the main        switch.    -   4. The second switch can turn on/off opposite of the main switch        in a complementary fashion. This creates a continuous current        mode condition, whenever it is desirable.    -   5. The second switch can be the same as the first switch.    -   6. Particularly in single switch Flyback circuits with the        synchronous rectification switch on the secondary side, the        secondary side switch can be used as the ZVS switch that turns        on ahead of the main switch.    -   7. In synchronous rectification mode, the second switch can turn        off when the inductor current reduces to zero. This may be        called discontinuous current condition mode. Then the second        switch stays off until right before the main switch turns on.    -   8. In situations when the voltage ringing across the main switch        is large enough to achieve ZVS, the second switch can remain off        and may not be activated.    -   9. During conditions such as light load, the second switch can        remain off without being activated.    -   10. This novel ZVS scheme applies to many circuit topologies        such as Buck, Buck/Boost, Boost, Flyback, Forward and other        converter topologies.    -   11. This novel ZVS scheme may be particularly useful for        circuits that use GaN as power devices.

Integration and Co-Packaging

Now referring to FIG. 14, in some embodiments one or more electroniccomponents may be integrated within a single electronic package 1400(i.e., co-packaged). In one embodiment MFET 1405 and SFET 1410 may beco-packaged in a unitary electronic package 1400. In one embodiment MFET1405 and SFET 1410 may each have external source, gate and drainconnections. An external connection may be an electrical connection thatis made outside of package 1400, such as a solder connection to anothercircuit board. In further embodiments, MFET 1405 may have external gateand source connections and an internal drain connection to a source ofSFET 1410, forming switch node (e.g., item 120 in FIG. 1). In someembodiments, the switch node may also have an external connection. Thedrain and gate of SFET 1410 may further have an external connections onpackage 1400. Co-packaging may result in reduced packaging cost,decreased size and improved performance for the power converter, asdiscussed in more detail below.

In further embodiments illustrated in FIG. 15, electronic package 1500may have a dual driver 1505 (i.e., a single die that drives both theMFET and the SFET) co-packaged with MFET 1405 and SFET 1410. In suchembodiments dual driver 1505 may be internally electrically connected tothe gate terminals of MFET 1405 and SFET 1410. Thus, gate driver 1505may have one or more external gate control connections on package 1500that are electrically coupled to the controller. Similar to package1400, package 1500 may also have external MFET source, switch node andSFET drain connections. In yet further embodiments, one or more otheractive or passive electronic components such as, but not limited tocapacitors, resistors, diodes and the like may be integrated within theunitary package. In other embodiments the controller (not shown) mayalso be co-packaged with driver 1505 and one or more of FETs 1405, 1410.In such embodiments, there may be one or more external controllerconnections along with the MFET and SFET connections discussed above.

In some embodiments, particularly in high frequency applications,co-packaging driver 1505 with one or more of FETs 1405, 1410 may enableimproved converter performance through the elimination of packaging andcomponent interconnect parasitics. All conductors and electricalcomponents possess parasitic elements. For instance, a resistor isdesigned to possess resistance, but will also possess unwanted parasiticcapacitance. Similarly, a conductor is designed to conduct an electricalsignal, but will also possess unwanted parasitic resistance andinductance. Parasitic elements cause propagation delays and impedancemismatches which limit the operating frequency of the converter. Thus,the elimination and or minimization of conductors and interconnectstructures between electronic components eliminates/minimizes parasiticelements that limit the maximum operating frequency of the converter.

In some embodiments electronic packages 1400, 1500 may be what are knownas organic multi-chip modules. An organic substrate 1450, such as, butnot limited to a printed circuit board, may be used as a mount for theFETs 1405, 1410 and/or driver 1505, controller and other components andmay also provide electrical interconnectivity between the devices withinthe package and/or between the devices and the system to which package1400, 1500 is mounted. In some embodiments one or more devices may beattached to the substrate with an electrically conductive material suchas, but not limited to, solder or epoxy. In some embodiments theelectronic devices may be electrically interconnected to the substratewith wire bonds while in further embodiments flip-chip devices,conductive columns or other electrical interconnects may be used. Anelectrically insulative potting compound 1455 may be molded on top ofthe substrate and around the electrical devices to provide environmentalprotection.

Interconnections between electronic packages 1400, 1500 and anotherelectronic system, such as a printed circuit board, may be referred toherein as external connections. External connections between electronicpackages 1400, 1500 and the system may be made with, but not limited to,solder or conductive epoxy. Other methods and structures may be used toco-package the devices without departing from the invention.

In further embodiments the package may be what is known as a lead-framebased multi-chip module where substrate 1450 is a metallic leadframe.Electrical interconnects between the devices and the leadframe may beperformed as discussed above. The one or more electronic devices may beattached to the leadframe and the assembly may be over molded withelectrically insulative potting compound 1455 as discussed above.External connections may also be formed as discussed above.

Now referring to FIG. 16 an example lead-frame layout 1600 for an S0-8,leadless chip carrier or other package is illustrated. MFET 1605 isplaced on first die paddle 1610 and SFET 1615 is placed on second diepaddle 1620. Connections may be made from MFET 1605 and SFET 1615 tofirst die paddle 1610 and/or peripheral connections 1625 using one ormore electrical conductors such as wirebonds, metallic straps, otherconductive interconnects. In such embodiments, the electronic packagemay have external connections for the gate, the source and the drain forMFET 1605 and SFET 1615, respectively. Alternatively, two of the FETconnections may form a switchnode connection, as discussed above. Infurther embodiments, driver circuit may be integrated on MFET 1605 orSFET 1615 die and thus the package would have one or more drivercontroller connections.

Now referring to FIG. 17 an example lead-frame layout 1700 for an SO-8,leadless chip carrier or other package is illustrated. In one embodimentdie 1705 may be an MFET and placed on first die paddle 1710 along withsecond die 1715 which may be an SFET. Connections may be made from firstdie 1705 and second die 1715 to die paddle 1710 and/or peripheralconnections 1725, as discussed above. The electronic package may haveone or more external electrical connections as discussed above. Infurther embodiments, first die 1705 may be a monolithically integrateddriver circuit along with an MFET and/or SFET device and die 1705 maycomprise a controller die. In such embodiments, the package may not haveexternal driver control connections and may have one or more controllerconnections such as, but not limited to driver power and feedbackconnections.

Now referring to FIG. 18, in some embodiments one or more electronicdevices may be fabricated on a monolithic semiconductor substrate 1800.More specifically, in one embodiment MFET region 1805 and SFET region1810 may both be fabricated on one semiconductor die 1820. In furtherembodiments, driver region 1815 may also be fabricated on singlesemiconductor die 1820 along with MFET region 1805 and SFET region 1810.Thus, in such embodiments, one die may contain the functionality ofthree or more single die. As discussed above, GaN FETs may beparticularly useful for such integration as they are lateral devices andregions 1805, 1810 and 1815 can all be electrically isolated from oneanother. In such embodiments, die 1800 may have one or more externalconnections such as, but not limited to source and drain connections forMFET region 1805 and SFET region 1810 along with one or more controllerconnections.

In yet further embodiments one or more of the semiconductor devices maybe manufactured on a substrate comprising gallium nitride (GaN). In oneembodiment one or more of the devices may be fabricated on a substratehaving a base of silicon with an epitaxially deposited layer of GaN. Inother embodiments different substrate configurations may be employed.

In some embodiments, GaN based devices may be particularly well suitedto switch at high frequencies due to their lower output capacitance, orCoss values. As discussed above, each time the FET turns on, the energystored in the output capacitance will be dissipated in the device. Asthe switching frequency increases, the power dissipation in the FET dueto discharging this energy increases proportionately, which may become alimiting factor in hard switching topologies.

In some embodiments, with regard to ZVS switched GaN devices, therelatively small Coss associated with GaN devices, on the order of 10picofarads, may enable faster discharging of the Coss and thus higherswitching frequencies. In further embodiments GaN devices also switchrelatively fast, on the order of 2 nanoseconds, enabling them to operateat high frequencies.

Further, in some embodiments GaN devices may be operated with relativelysmall driver circuits, even at high voltages, making the size and thecost of the driver circuit attractive for high voltage applications. Yetfurther, since GaN devices are lateral, as discussed above, and thedriver circuit may be relatively small, some embodiments may benefitfrom integrating the driver circuit monolithically or co-packaged withone or more of the FETs, as discussed in more detail below.

Integrated Driver Circuits

In some embodiments, it may be beneficial to integrate a driver circuiton the same die as, or co-packaged with either of the FETs to minimizeinterconnect parasitics, enabling higher switching frequencies, asdiscussed above. Integrating more than two circuit elements on a singledie may eliminate even more parasitic elements and further increase theoperating frequency as well as reduce costs.

The schematic in FIG. 19 illustrates a two switch half-bridge Flybackcircuit 1900 that incorporates some embodiments of the invention. Eachhigh side or low side FET has an integrated driver circuit, also calleda pre-driver to turn on or off according to gate signals (GateH andGateL). In some embodiments the gate control can be one signal while inother embodiments it may be two separate signals. In one embodiment,GateH may be used to control the high side FET and GateL may be used tocontrol the low side FET. In further embodiments the high side circuit(see dashed box in FIG. 19) can move up and down by approximately 600V.Therefore, in this embodiment the gate drive circuit for the high sideFET may float with V_(SW) node. The gate drive control signal for thehigh side may be translated through a level shifter circuit, which maybe integrated into the half bridge block. As defined herein, a levelshifter circuit may convert the voltage potential of a signal to adifferent voltage potential.

In further embodiments, a startup circuit may be included to deliverbias current to the half bridge block during initial power up. Thestartup circuit may draw current from the high voltage V+ node. Once thecontrol circuit is fully biased, the startup FET can be turned off toreduce power loss. In yet further embodiments, the high side circuit mayneed a bias supply to drive the main FET. In some embodiments abootstrap circuit may be used to deliver power to high side circuit todrive the main FET. In one embodiment, the bootstrap circuit can be usedto charge a high side capacitor when the low side FET turns on.

In some embodiments, two switch half-bridge Flyback circuit 1900 may beimplemented as a multi-chip hybrid solution including GaN-based highside and GaN-based low side switches. In one embodiment, the entirecircuit may be disposed on a single die. In further embodiments the highside circuit (see dashed box in FIG. 19) may be on one die and theremainder of the circuit may be on another die. In yet furtherembodiments both the high side power switch and the low side powerswitch may be discrete GaN-based power switches and may be co-packagedwith a half-bridge driver to form a half-bridge power block.

Now referring to FIG. 20, each GaN transistor may have its own drivecircuit monolithically integrated with the FET. Circuit 2000 illustratesan embodiment of a simplified integrated drive circuit for a single FET.For example, MFET 105 and SFET 106 (see FIG. 1) may each contain acircuit similar to circuit 2000. In some embodiments the main FET can bea high voltage FET. In one embodiment, an internal low voltage pull-downFET can be used to hold the gate of main FET low while minimizing layoutrelated noise injection so the main FET doesn't turn on accidentally. Inthis simplified integrated drive circuit, two gate signals may beavailable. In one embodiment, the main FET gate may control the main FETand the pull-down FET gate may control the small FET. The timing of bothgate signals may be handled by an external control circuit. In furtherembodiments, the gate terminals may be ESD (electrical static discharge)sensitive. In one embodiment integrated clamp circuits may be used toprotect the gate from ESD voltage.

Now referring to FIG. 21 an integrated drive circuit 2100 for a singleFET is illustrated. Circuit 2100 may be used as a building block formore complicated multi-switch circuits such as two switch half bridgeFlyback circuit 1900 illustrated in FIG. 19. Continuing to refer to FIG.19, in some embodiments each pre-driver and each power FET may use anintegrated drive circuit 2100. In contrast to circuit 2000 illustratedin FIG. 20, circuit 2100 has both integrated pull-up and integratedpull-down transistors. In one embodiment a single gate control signalmay be used that may control the pull-up and pull-down using properlogic. In further embodiments, the DC biased voltage node may providedrive current to the main gate. In other embodiments, a charge pumpcircuit may be used to boost voltage internally to drive the pull-upFET. In further embodiments, an ESD structure may be integrated toprotect the input signal from high ESD voltages. In yet furtherembodiments, an additional FET and diode may be used to provide properlogic and timing control.

In some embodiments, an advantage of an integrated drive circuit maymake the device relatively easy for a circuit designer to incorporateand it may also provide a rugged circuit with integrated protection.Further, in some embodiments, the input signal may be a logic signalthat doesn't need to be strong to provide gate drive current. Therefore,such circuits may save the circuit designer from adding an externaldrive device. Moreover, in some embodiments noise may be suppressedinternally, and noise coupling from an external circuit may beminimized.

In the foregoing specification, embodiments of the invention have beendescribed with reference to numerous specific details that may vary fromimplementation to implementation. The specification and drawings are,accordingly, to be regarded in an illustrative rather than a restrictivesense. The sole and exclusive indicator of the scope of the invention,and what is intended by the applicants to be the scope of the invention,is the literal and equivalent scope of the set of claims that issue fromthis application, in the specific form in which such claims issue,including any subsequent correction.

1. A power conversion circuit comprising: a voltage source having firstand second output terminals; a first solid-state bidirectional switchhaving a pair of first power terminals and a first gate terminal, thepair of first power terminals connected between the first outputterminal and a switch node; a first driver circuit transmitting a firstgate drive signal to the first gate terminal in response to receiving afirst gate control signal; a second solid-state bidirectional switchhaving a pair of second power terminals and a second gate terminal, thepair of second power terminals connected between the switch node and thesecond output terminal; a second driver circuit transmitting a secondgate drive signal to the second gate terminal in response to receiving asecond gate control signal; and a controller transmitting the first gatecontrol signal to create an oscillating voltage at the switch node andtransmitting the second gate control signal based on the oscillatingvoltage such that the second switch turns on before the first switchwith a duration that is less than a time that the first switch is off.2. The power conversion circuit of claim 1 wherein the first and secondsolid-state bidirectional switches comprise gallium nitride.
 3. Thepower conversion circuit of claim 2 wherein the first and secondsolid-state bidirectional switches operate at frequencies between 50 kHzand 100 MHz.
 4. The power conversion circuit of claim 2 wherein thefirst and second solid-state bidirectional switches operate at voltagesbetween 100 volts and 600 volts.
 5. The power conversion circuit ofclaim 2 wherein the first and the second driver circuits are co-packagedwith the first and second solid-state bidirectional switches.
 6. Thepower conversion circuit of claim 2 wherein the first and secondsolid-state bidirectional switches are co-packaged in a unitaryelectronic package.
 7. The power conversion circuit of claim 1 whereinthe first and second solid-state bidirectional switches are disposed ona monolithic die comprising gallium nitride.
 8. The power conversioncircuit of claim 1 wherein the first switch driver circuit and the firstsolid-state bidirectional switch are monolithically integrated on afirst die, and the second switch driver circuit and the secondsolid-state bidirectional switch are monolithically integrated on asecond die.
 9. The power conversion circuit of claim 8 wherein the firstdie and the second die are co-packaged in a unitary electronic package.10. The power conversion circuit of claim 1 wherein a level shifter isdisposed between the first and the second driver circuits.
 11. Anelectronic power conversion component comprising: a substrate; a firstdie secured to the substrate and comprising a first solid-statebidirectional switch having a pair of first power terminals and a firstgate terminal wherein the pair of first power terminals are connectedbetween a first external connection on the component and a secondexternal connection on the component; a second die secured to thesubstrate and comprising a second solid-state bidirectional switchhaving a pair of second power terminals and a second gate terminalwherein the pair of second power terminals are connected between thesecond external connection on the component and a third externalconnection on the component; and an electrically insulative moldcompound encapsulating a top surface of the substrate and the first andthe second solid-state bidirectional switches.
 12. The power conversioncomponent of claim 11 further comprising an overmolded printed circuitboard.
 13. The power conversion component of claim 11 further comprisingan overmolded leadframe.
 14. The power conversion component of claim 11wherein the first and second die comprise gallium nitride.
 15. The powerconversion component of claim 11 wherein a first switch driver circuitand the first solid-state bidirectional switch are monolithicallyintegrated on the first die, and a second switch driver circuit and thesecond solid-state bidirectional switch are monolithically integrated onthe second die.
 16. The power conversion component of claim 15 furthercomprising a first gate control connection electrically coupled to thefirst switch driver and forming a fourth external connection on thecomponent, and a second gate control connection electrically coupled tothe second switch driver and forming a fifth external connection on thecomponent.
 17. A method of operating a power conversion circuit, themethod comprising: supplying power to the circuit with a voltage sourcehaving a first and a second output terminal; transmitting a first gatecontrol signal to a first driver circuit, in response the first drivercircuit transmitting a first gate drive signal to a gate of a firstsolid-state bidirectional switch, the first switch having a pair offirst power terminals connected between the first output terminal and aswitch node; transmitting a second gate control signal to a seconddriver circuit, in response the second driver circuit transmitting asecond gate drive signal to a gate of a second solid-state bidirectionalswitch, the second switch having a pair of second power terminalsconnected between the second output terminal and the switch node; andoperating a controller that transmits the first gate control signal tocreate an oscillating voltage at the switch node and transmits thesecond gate control signal based on the oscillating voltage such thatthe second switch remains in an on state for a duration that is lessthan a time that the first switch is in an off state.
 18. The method ofclaim 17 wherein a capacitor is connected in series between the secondswitch and the second output terminal and allows a reversal of adirection of current in the circuit when the second switch istransitioned to an off state causing an output capacitance of the firstswitch to discharge to approximately 0 volts.
 19. The method of claim 17wherein the first and second switches operate at frequencies between 50kHz and 100 MHz and voltages between 100 volts and 600 volts.
 20. Themethod of claim 17 wherein the first and second solid-statebidirectional switches comprise gallium nitride and are co-packagedwithin a unitary electronic component